Transcript ECEN5817L40

Zero-voltage transition converters
The phase-shifted full bridge converter
Buck-derived full-bridge converter
A popular converter for server frontend power systems
Zero-voltage switching of each halfbridge section
Each half-bridge produces a square
wave voltage. Phase-shifted control of
converter output
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Efficiencies of 90% to 95% regularly
attained
Controller chips available
Lectures 39-40
Actual
waveforms,
including
resonant
transitions
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Issues with this converter
It’s a good converter for many applications requiring isolation. But…
1. Secondary-side diodes operate with zero-current switching. They
require snubbing or other protection to avoid failure associated with
avalanche breakdown
2. The resonant transitions reduce the effective duty cycle and conversion
ratio. To compensate, the transformer turns ratio must be increased,
leading to increased reflected load current in the primary-side elements
3. During the D’Ts interval when both output diodes conduct, inductor Lc
stores energy (needed for ZVS to initiate the next DTs interval) and its
current circulates around the primary-side elements—causing
conduction loss
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Lectures 39-40
Result of analysis
Basic configuration: full bridge ZVT
• Phase shift  assumes the role of duty cycle d in converter equations
• Effective duty cycle is reduced by the resonant transition intervals
• Reduction in effective duty cycle can be expressed as a function of the
form FPZVT(J), where PZVT(J) is a negative number similar in magnitude
to 1. F is generally pretty small, so that the resonant transitions do not
require a substantial fraction of the switching period
• Circuit looks symmetrical, but the control, and hence the operation,
isn’t. One side of bridge loses ZVS before the other.
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Effect of ZVT: reduction of effective duty cycle
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Phase-shifted control
Approximate waveforms
and results
(as predicted by
analysis of the
parent hardswitched converter)
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Diode switching
analysis
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Diode commutation: intervals 3 and 4
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Waveforms: ZCS
of D6
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Approaches to snub the diode ringing
(a) conventional diode snubber
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Approaches to snub the diode ringing
(b) conventional passive voltage-clamp snubber
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Approaches to snub the diode ringing
(c) simplify to one passive voltage-clamp snubber
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Approaches to snub the diode ringing
(d) improvement of efficiency in voltage-clamp snubber
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Approaches to snub the diode ringing
(e) active clamp lossless snubber
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Approaches to snub the diode ringing
(f) primary-side lossless voltage clamp
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Another application of the ZVT: DC Transformer
Operate at a fixed conversion ratio with high duty cycle, leading to
high efficiency—avoids the problems of circulating currents
Use other elements in the system to regulate voltage
PFC
350 V
ZVT
5V
DC-DC
1V
Load
AC
line
isolation
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DC-DC
Load
DC-DC
Load
Lectures 39-40
Active clamp circuits
Can be viewed as a lossless voltage-clamp snubber that employs a
current-bidirectional switch
See Vinciarelli patent (1982) for use in forward converter
Related to other half-bridge ZVS circuits
Can be added to the transistor in any PWM converter
Not only adds ZVS to forward converter, but also resets transformer better,
leading to better transistor utilization than conventional reset circuit
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Lectures 39-40
The conventional forward converter
• Max vds = 2Vg + ringing
• Limited to D < 0.5
• On-state transistor current is P/DVg
• Magnetizing current must operate in DCM
• Peak transistor voltage occurs during
transformer reset
• Could reset the transformer with less voltage
if interval 3 were reduced
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The active-clamp forward converter
• Better transistor/transformer
utilization
• ZVS
• Not limited to D < 0.5
Transistors are driven in usual half-bridge manner:
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Approximate analysis:
ignore resonant transitions, dead times, and resonant elements
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Charge balance
Vb can be viewed as a flyback converter output. By use of a current-bidirectional switch,
there is no DCM, and LM operates in CCM.
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Peak transistor voltage
Max vds = Vg + Vb = Vg /D’
which is less than the conventional value of 2 Vg when D > 0.5
This can be used to considerable advantage in practical applications where
there is a specified range of Vg
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Design example
270 V ≤ Vg ≤ 350 V
max Pload = P = 200 W
Compare designs using conventional 1:1 reset winding and using active
clamp circuit
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Lectures 39-40
Conventional case
Peak vds = 2Vg + ringing
= 700 V + ringing
Let’s let max D = 0.5 (at Vg = 270 V),
which is optimistic
Then min D (at Vg = 350 V) is
(0.5)(270)/(350) = 0.3857
The on-state transistor current, neglecting ripple, is given by
 ig  = DnI = Did-on
with P = 200 W = Vg  ig  = DVg id-on
So id-on = P/DVg = (200W) / (0.5)(270 V) = 1.5 A
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Lectures 39-40
Active clamp case:
scenario #1
Suppose we choose the same turns ratio as in the conventional design.
Then the converter operates with the same range of duty cycles, and
the on-state transistor current is the same. But the transistor voltage is
equal to Vg / D’, and is reduced:
At Vg = 270 V:
D = 0.5
peak vds = 540 V
At Vg = 350 V:
D = 0.3857
peak vds = 570 V
which is considerably less than 700 V
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Lectures 39-40
Active clamp case:
scenario #2
Suppose we operate at a higher duty cycle, say, D = 0.5 at Vg = 350 V.
Then the transistor voltage is equal to Vg / D’, and is similar to the
conventional design under worst-case conditions:
At Vg = 270 V:
At Vg = 350 V:
D = 0.648
D = 0.5
peak vds = 767 V
peak vds = 700 V
But we can use a lower turns ratio that leads to lower reflected current in
Q1:
id-on = P/DVg = (200W) / (0.5)(350 V) = 1.15 A
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Lectures 39-40