Transcript Ground and Power Planes

Ground and Power Planes
Chris Allen ([email protected])
Course website URL
people.eecs.ku.edu/~callen/713/EECS713.htm
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Ground planes
Ground planes were assumed in previous discussions
discussion on vias
layer stack up (MCMs)
microstrip and stripline transmission line
Ground planes serve several functions
return path for signal current
stable voltage reference
Focus here will be primarily on return path and crosstalk
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Ground planes
Ground planes as current return path
DC current flows along path of least resistance
Consider the circuit 
for I0 = 1 A
I1 = I0 1000/1001 = 0.999 A
I2 = 1 mA
for AC (f > 0) current follows the path of
least impedance
Z = R + jX
XL = L , XC = -1/C
As discussed previously, inductance compels
the return path current to flow beneath
the signal trace (proximity effect)
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Ground planes
For microstrip transmission lines
at frequencies where L >> R
the distribution of return current follows
ID  
Io
1

, A/m
2
H 1  D H 
small H, the current is more concentrated
large H, the current is less concentrated
For striplines
the current follows a similar
distribution based on how
the current is split between the
two return paths
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Ground planes
Return-path current and crosstalk
Adjacent transmission lines can experience crosstalk as the current
(and resulting magnetic field) induces a signal in the passive
(inactive) trace
The induced crosstalk falls off with the square of the increasing
distance
crosstalk 
K
2
1  D H 
Where
K = f (Tr, trace length) < 1
K can be determined experimentally
Unless K is known, we will assume K = 1 to be conservative
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Ground planes
Return-path current and crosstalk
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Ground planes
Return-path current and crosstalk
Interruptions in the ground plane that prevent electrical continuity
disrupt current flow effectively lengthening the return path
This increase in circuit loop area increases the path inductance by L
L  5 D ln D W  , nH
D is the slot length (the perpendicular extent of the current diversion away from
the signal trace) in inches
W is the trace width, in inches
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Ground planes
Effects of ground-plane interruptions
Increases signal rise time
Similar to L-R circuit
L
Tr L R   2.2
 Tr total  Tr2L R   Tr2
2 Zo
assumes resistance on either side of inductance is Zo
Increases crosstalk due to current coupling to nearby traces
Clearly breaks in the ground plane cause problems
So why would designer break ground plane continuity?
Perhaps to route signals that won’t ‘fit’ on other routing layers
Note: the phenomenon of breaks in the ground plane increasing path inductance is
used in non-destructive tests to locate metal fatigue wherein micro-cracks disrupt
surface current flow resulting in magnetic fields uncharacteristic of solid conductors
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Ground planes
Effects of ground-plane interruptions
Example – slotted ground plane
10G GaAs technology, Tr = 150 ps
Zo = 50 
Trace width, 10 mils (0.010”)
LnH  5 D ln D W   L  4 nH
Tr due to slot, Tr L R   2.2
D = 0.25”
L
 88 ps
2 Zo
total Tr, Tr total  Tr2L R   Tr2 
88 ps2  150 ps2
 174 ps
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Ground planes
Examples of discontinuous ground plane
Ground plane grid scheme
Saves board area at expense of mutual inductance
Power dist on top layer / GND on bottom
Fine for low-speed CMOS and TTL
High-freq return-path current
uses bypass capacitors
to stay near signal current
on way back to drive gate
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Ground planes
Examples of discontinuous ground plane
Ground fingers layout
Returning signal currents must go all the way around the board edge
resulting in massive amounts of self-inductance and mutual inductance
Power and ground wiring on single layer
Found in prototyping circuit boards
Works well with low-speed logic
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Ground planes
Grounded guard traces
Guard traces between signal traces may reduce crosstalk
Coupling between adjacent microstrip line can be halved, if sufficient
space is available, by inserting between them a third trace (grounded at
both ends), called a guard trace,.
Coupling can be halved again if this guard trace is connected to the ground
plane at frequent intervals.
However the best way to manage crosstalk is provide adequate
separation and provide continuous ground plane
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Ground planes
Near-end and far-end crosstalk
The coupled signal on the passive trace has different characteristics
when viewed from the near end vs. the far end
continuous coupling or
discrete coupling regions
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Ground planes
Near-end and far-end crosstalk
First the inductive crosstalk
Consider a positive-going step
Transient induces current in ckt C-D
Forward-going negative ‘blips’ prop at
same velocity as driving signal so they
add constructively (similar to how shock waves add to form sonic boom)
Reverse-going positive ‘blips’ have
distributed start times and travel times
 blips blend into one long, low blob
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Ground planes
Near-end and far-end crosstalk
Now the capacitive crosstalk
Again a positive-going step
Transient couples current in ckt C-D
Forward-going positive ‘blips’ prop at
same velocity as driving signal
add constructively
Again, reverse-going positive ‘blips’ have
distributed start times and travel times
 blips blend into one long, low blob
Note the positive polarity of
the capacitively-coupled blips
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Ground planes
Near-end and far-end crosstalk
Normally the inductive and capacitive crosstalk voltages are of
comparable magnitude
The forward crosstalk components should therefore cancel at the far end
The reverse crosstalk components reinforce at the near end
Stripline circuits are typically well balanced between inductive and
capacitive coupling while microstrip lines generally have slightly less
capacitive coupling than inductive
Reverse-coupled signals can
become an issue if the nearend impedance mismatch
is significant
Example shows reversecoupled signal reflected
from a low-impedance
(Rs < Zo    –1) driver
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Ground planes
Near-end and far-end crosstalk
Example – coupled 50- microstrip lines
w = 15 mils, h = 10 mils, D = 45 mils, ℓ = 1.5”
Crosstalk 
1
1  D H 
2
 0.047
for GaAs technology, V = 1.5 V, Tr = 150 ps
crosstalk  V = 71 mV
For delay = 180 ps/in, Tp = 270 ps
Vc = Area/2Tp
= 10.6x10-12/540 ps
= 19.6 mV
Capacitive crosstalk
effectively cancels blip at D
doubles crosstalk at C  39 mV
if source impedance, Rs = 8 
s = -0.72, at 3Tp (810 ps), -28 mV (1.9% of V)
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Power planes
Power planes
Good idea to dedicate a power plane for each supply voltage that is
used widely
example: ECL designs VEE = -4.7 V and VTT = -2 V
Substantial currents pass through these planes (several amps)
1 oz. or 2 oz. copper should be used
Desired properties: low resistance, low inductance
To connect components to power, simply sink a via to the appropriate
power plane
One power plane per voltage is generally adequate
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Board stack-up of layers
Designing a stack-up of layers
N: number of conducting layers
A double-sided board has N = 2
A multilayer board has N>2 layers
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Board stack-up of layers
Multilayer circuit board
Normally vias penetrate entire stack
Blind or buried vias may also be used
Blind vias begin on one surface but do not
penetrate the board thickness
Buried vias begin and end on internal layers
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Board stack-up of layers
Multilayer circuit board
Routing issues
Components are mounted on outer layers (exceptions are buried components)
Surface-mounted components don’t require vias for each device pin
Permits higher density component placement and signal routing
Through-hole components require vias for component leads
these vias obstruct routing paths on internal and backside layers
Thermal vias beneath components also obstruct routing paths on internal
and backside layers
When routing signals on the 2 outer layers, components obstruct routing,
often requiring internal routing layers
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Board stack-up of layers
Example multilayer circuit board stack-ups
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Board stack-up of layers
Example multilayer circuit board stack-ups
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Board stack-up of layers
Example multilayer circuit board stack-ups
To minimize warping, it is desirable for the stack-up to be symmetrical
(this example is symmetric)
Thinner copper layers enable more
precise stripline trace widths
Thicker power planes provide
lower resistance
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Signal routing layers
Routing layers
Signal routing on layers are usually divided into x (east-west) and
y (north-south) trace directions
Maximizes utilization of limited routing space
Lower speed signals can share adjacent layers (since crosstalk is not
an issue)
For high-speed signal routing, layout rules must be specified
Trace width, W
Minimum trace separation, S, (assumes r, and b or h are known)
Metal thickness, t, from copper weight (1/2 oz. or 1 oz.)
Ground and power planes may use 1 oz. or 2 oz. copper for lower
resistance
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Signal routing layers
Routing layers
How many routing layers are needed?
Observed relationship: Rent’s rule
p avg 
for a board width, X
board height, Y
average trace pitch, pavg
number of signals (connections), N
number of routing layers, M
XY
N
2.7 M
pitch = D = S + W
X, Y, and N known from system constraints
W, D determined from Zo and crosstalk analysis
 minimum pavg is known
Leaves M (number of routing layers) as the unknown
Example, W = 15 mils, S = 45 mils  pavg = 0.06
N = 150, X = 3”, Y = 3”  M = 1.1  = 2
Rent’s rule says at least 2 routing layers needed
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