Transcript ch06

Digital Design
Chapter 6:
Optimizations and Tradeoffs
Slides to accompany the textbook Digital Design, First Edition,
by Frank Vahid, John Wiley and Sons Publishers, 2007.
http://www.ddvahid.com
Copyright © 2007 Frank Vahid
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6.1
Introduction
• We now know how to build digital circuits
– How can we build better circuits?
• Let’s consider two important design criteria
– Delay – the time from inputs changing to new correct stable output
– Size – the number of transistors
– For quick estimation, assume
Transforming F1 to F2 represents
an optimization: Better in all
criteria of interest
• Every gate has delay of “1 gate-delay”
• Every gate input requires 2 transistors
• Ignore inverters
w
x
y
F1
w
x
y
4 transistors
1 gate-delay
w
x
F1 = wxy + wxy’
(a)
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Frank Vahid
= wx(y+y’) = wx
F2
F2 = wx
(b)
20
size
(transistors)
16 transistors
2 gate-delays
si
F1
15
e
t 10rs)
zo
5
r(t
F2
1 2 3 4
delay (gate-delays)
(c)
2
Note: Slides with animation are denoted with a small red "a" near the animated items
Introduction
• Tradeoff
– Improves some, but worsens other, criteria of interest
Transforming G1 to G2
represents a tradeoff: Some
criteria better, others worse.
G1
w
y
z
G1 = wx + wy + z
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w
20
12 transistors
3 gate-delays
x
y
z
G2
G2 = w(x+y) + z
e
z
size
(transistors)
w
x
14 transistors
2 gate-delays
15
10
G1
G2
5
1 2 3 4
delay (gate-delays)
3
Introduction
Tradeoffs
size
All criteria of interest
are improved (or at
least kept the same)
size
Optimizations
e
z si
delay
Some criteria of interest
are improved, while
others are worsened
e
zsi
delay
siansis
• We obviously prefer optimizations, but often must accept
tradeoffs
e
– You can’t build a car that is the most comfortable,
and has the best
z
fuel efficiency, and is the fastest – you have to give up something to
gain other things.
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4
Combinational Logic Optimization and Tradeoffs
• Two-level size optimization using
algebraic methods
Example
F = xyz + xyz’ + x’y’z’ + x’y’z
F = xy(z + z’) + x’y’(z + z’)
– Goal: circuit with only two levels (ORed
AND gates), with minimum transistors
F = xy*1 + x’y’*1
• Though transistors getting cheaper
(Moore’s Law), they still cost something
F = xy + x’y’
• Define problem algebraically
x
F
y’
6 gate inputs =
12 transistors
n
0
0
0
m
y
x
y’
x’
m
n
n
y
F
y’
n
m
x
1
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m
x’
• F = abc + abc’ is sum-of-products; G =
w(xy + z) is not.
• Each literal and term translates to a
gate input, each of which translates to
about 2 transistors (see Ch. 2)
• Ignore inverters for simplicity
4 literals + 2
terms = 6
gate inputs
y
– Sum-of-products yields two levels
– Transform sum-of-products equation to
have fewest literals and terms
6.2
x’
Note: Assuming 4-transistor 2-input AND/OR circuits;
in reality, only NAND/NOR are so efficient.
1
1
5
Algebraic Two-Level Size Minimization
• Previous example showed common
algebraic minimization method
– (Multiply out to sum-of-products, then)
– Apply following as much possible
• ab + ab’ = a(b + b’) = a*1 = a
• “Combining terms to eliminate a variable”
– (Formally called the “Uniting theorem”)
– Duplicating a term sometimes helps
• Note that doesn’t change function
– c + d = c + d + d = c + d + d + d + d ...
F = xyz + xyz’ + x’y’z’ + x’y’z
F = xy(z + z’) + x’y’(z + z’)
a
F = xy*1 + x’y’*1
F = xy + x’y’
F = x’y’z’ + x’y’z + x’yz
F = x’y’z’ + x’y’z + x’y’z + x’yz
F = x’y’(z+z’) + x’z(y’+y)
a
F = x’y’ + x’z
– Sometimes after combining terms, can
combine resulting terms G = xy’z’ + xy’z + xyz + xyz’
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G = xy’(z’+z) + xy(z+z’)
G = xy’ + xy (now do again)
G = x(y’+y)
G=x
a
6
Karnaugh Maps for Two-Level Size Minimization
F yz
• Easy to miss “seeing” possible opportunities
to combine terms
• Karnaugh Maps (K-maps)
x
– Graphical method to help us find
opportunities to combine terms
– Minterms differing in one variable are adjacent
in the map
– Can clearly see opportunities to combine
terms – look for adjacent 1s
• For F, clearly two opportunities
• Top left circle is shorthand for x’y’z’+x’y’z =
x’y’(z’+z) = x’y’(1) = x’y’
• Draw circle, write term that has all the literals
except the one that changes in the circle
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Notice not in binary order
01
11
10
0
x’y’z’ x’y’z
x’yz x’yz’
1
xy’z’
xyz
xy’z
a
xyz’
F = x’y’z + xyz + xyz’ + x’y’z’
F
yz
x
00
01
11
10
0
1
1
0
0
1
0
0
1
1
a
yz
x
00
01
11
10
0
1
1
0
0
1
0
0
1
1
F = x’y’ + xy x’y’
Easier than all that algebra:
K-map
Treat left & right as adjacent too
F
– Circle xy, x=1 & y=1 in both cells of the circle, but
z changes (z=1 in one cell, 0 in the other)
• Minimized function: OR the final terms
00
F = xyz + xyz’ + x’y’z’ + x’y’z
F = xy(z + z’) + x’y’(z + z’)
F = xy*1 + x’y’*1
F = xy + x’y’
a
xy
7
K-maps
• Four adjacent 1s means
two variables can be
eliminated
– Makes intuitive sense – those
two variables appear in all
combinations, so one must be
true
– Draw one big circle –
shorthand for the algebraic
transformations above
G = xy’z’ + xy’z + xyz + xyz’
G = x(y’z’+ y’z + yz + yz’) (must be true)
G = x(y’(z’+z) + y(z+z’))
G = x(y’+y)
G=x
G
yz
x
00
01
11
10
0
0
0
0
0
1
1
1
1
1
x
G
yz
x
Draw the biggest
0
circle possible, or
you’ll have more terms
1
than really needed
xy’
Digital Design
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Frank Vahid
00
01
11
10
0
0
0
0
1
1
1
1
xy
8
K-maps
• Four adjacent cells can be in
shape of a square
• OK to cover a 1 twice
H
H = x’y’z + x’yz + xy’z + xyz
yz
x
00
01
0
1
0
– Just like duplicating a term
(xy appears in all combinations)
11
10
1
0
a
• Remember, c + d = c + d + d
1
• No need to cover 1s more than
once
0
I
– Yields extra terms – not minimized
1
1
0
yz
x
0
z
y’z
00
01
11
10
0
1
0
0
a
J
yz
x
x’y’
1
y’z
1
1
1
x
00
01
11
10
0
1
1
0
0
1
0
1
1
0
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1
xz
a
The two circles are shorthand for:
I = x’y’z + xy’z’ + xy’z + xyz + xyz’
I = x’y’z + xy’z + xy’z’ + xy’z + xyz + xyz’
I = (x’y’z + xy’z) + (xy’z’ + xy’z + xyz + xyz’)
I = (y’z) + (x)
9
K-maps
• Circles can cross left/right sides
K
x
– Remember, edges are adjacent
• Minterms differ in one variable only
• Circles must have 1, 2, 4, or 8
cells – 3, 5, or 7 not allowed
– 3/5/7 doesn’t correspond to
algebraic transformations that
combine terms to eliminate a
variable
• Circling all the cells is OK
– Function just equals 1
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x’y’z
yz
00
01
11
10
0
0
1
0
0
1
1
0
0
1
00
01
11
10
0
0
0
0
0
1
1
1
1
0
00
01
11
10
0
1
1
1
1
1
1
1
1
1
L
xz’
yz
x
E
yz
x
1
10
K-maps for Four Variables
F
• Four-variable K-map follows
same principle
wx
w’xy’
– Adjacent cells differ in one
variable
– Left/right adjacent
– Top/bottom also adjacent
• 5 and 6 variable maps exist
00
01
11
10
00
0
0
1
0
01
1
1
1
0
11
0
0
1
0
10
0
0
1
0
yz
G
– But hard to use
– But not very useful – easy to do
algebraically by hand F z
y
0
0
1
F=w’xy’+yz
yz
wx
• Two-variable maps exist
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yz
00
01
11
10
00
0
1
1
0
01
0
1
1
0
11
0
1
1
0
10
0
1
1
0
1
z
G=z
11
Two-Level Size Minimization Using K-maps
General K-map method
1. Convert the function’s equation into
sum-of-products form
2. Place 1s in the appropriate K-map
cells for each term
3. Cover all 1s by drawing the fewest
largest circles, with every 1
included at least once; write the
corresponding term for each circle
4. OR all the resulting terms to create
the minimized function.
Example: Minimize:
G = a + a’b’c’ + b*(c’ + bc’)
1. Convert to sum-of-products
G = a + a’b’c’ + bc’ + bc’
2. Place 1s in appropriate cells
G
bc
a
a’b’c’
0
00
01
11
10
1
0
0
1
a
1
1
1
1
1
a
3. Cover 1s
G
bc
a
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bc’
00
01
11
10
0
1
0
0
1
1
1
1
1
1
4. OR terms: G = a + c’
c’
a
12
Two-Level Size Minimization Using K-maps
– Four Variable Example
• Minimize:
a’b’c’d’
a’b’cd’
ab’c’d’ a’bd
a’bcd’
– H = a’b’(cd’ + c’d’) + ab’c’d’ + ab’cd’
+ a’bd + a’bcd’
1. Convert to sum-of-products:
– H = a’b’cd’ + a’b’c’d’ + ab’c’d’ +
ab’cd’ + a’bd + a’bcd’
2. Place 1s in K-map cells
3. Cover 1s
4. OR resulting terms
H
ab
00
01
11
10
00
1
0
0
1
01
0
1
1
1
a’bc
11
0
0
0
0
a’bd
10
1
0
0
1
H = b’d’ + a’bc + a’bd
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ab’cd’
cd
b’d’
a
Funny-looking circle, but
remember that left/right
adjacent, and top/bottom
adjacent
13
Don’t Care Input Combinations
• What if particular input combinations
can never occur?
– e.g., Minimize F = xy’z’, given that
x’y’z’ (xyz=000) can never be true,
and that xy’z (xyz=101) can never be
true
– So it doesn’t matter what F outputs
when x’y’z’ or xy’z is true, because
those cases will never occur
– Thus, make F be 1 or 0 for those
cases in a way that best minimizes
the equation
• On K-map
– Draw Xs for don’t care combinations
• Include X in circle ONLY if minimizes
equation
Digital
Design include other Xs
• Don’t
Copyright © 2006
Frank Vahid
F
y’z’
yz
x
00
01
11
10
0
X
0
0
0
1
1
X
0
0
Good use of don’t cares
F
yz
x
y’z’
unneeded
00
01
11
10
0
X
0
0
0
1
1
X
0
0
xy’
Unnecessary use of don’t
cares; results in extra term
14
Minimizization Example using Don’t Cares
• Minimize:
– F = a’bc’ + abc’ + a’b’c
– Given don’t cares: a’bc, abc
F
bc
a
0
a’c
b
00
01
11
10
0
1
X
1
a
• Note: Use don’t cares with
caution
– Must be sure that we really don’t
care what the function outputs for
that input combination
– If we do care, even the slightest,
then it’s probably safer to set the
output to 0
Digital Design
Copyright © 2006
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1
0
0
X
1
F = a’c + b
15
Minimization with Don’t Cares Example:
Sliding Switch
• Switch with 5 positions
1
– 3-bit value gives position in
binary
2
3
4
5
x
• Want circuit that
– Outputs 1 when switch is in
position 2, 3, or 4
– Outputs 0 when switch is in
position 1 or 5
– Note that the 3-bit input can
never output binary 0, 6, or 7
• Treat as don’t care input
combinations
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2,3,4,
detector
y
z
G
yz
Withou
x
t don’t
cares:
0
F = x’y
1
+ xy’z’
G
00
01
11
10
0
0
1
1
x’y
a
xy’z’
1
0
yz
x
G
0
0
y
00
01
11
10
0
X
0
1
1
1
1
0
X
X
a
With don’t
cares:
z’ F = y + z’
16
Automating Two-Level Logic Size Minimization
• Minimizing by hand
– Is hard for functions with 5 or
more variables
– May not yield minimum cover
depending on order we choose
– Is error prone
• Minimization thus typically
done by automated tools
– Exact algorithm: finds optimal
solution
– Heuristic: finds good solution,
but not necessarily optimal
I
yz
x
0
01
11
10
1
1
1
0
(a)
a
1
1
y’z’
I
x
0
0
x’y’
1
1
yz
xy
4 terms
yz
00
01
11
10
1
1
1
0
a
(b)
1
y’z’
Digital Design
Copyright © 2006
Frank Vahid
00
1
0
x’z
1
1
xy
Only 3 terms
17
Basic Concepts Underlying Automated Two-Level
Logic Minimization
• Definitions
F yz
x
00
– On-set: All minterms that define
when F=1
– Off-set: All minterms that define
when F=0
– Implicant: Any product term
(minterm or other) that when 1
causes F=1
• On K-map, any legal (but not
necessarily largest) circle
• Cover: Implicant xy covers
minterms xyz and xyz’
– Expanding a term: removing a
variable (like larger K-map circle)
• xyz  xy is an expansion of xyz
Digital Design
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Frank Vahid
x’y’z
01 11
10
0
0
1
0
0
1
0
0
1
1
xyz
xyz’
xy
4 implicants of F
Note: We use K-maps here just for
intuitive illustration of concepts;
automated tools do not use K-maps.
•
Prime implicant: Maximally
expanded implicant – any
expansion would cover 1s not in
on-set
• x’y’z, and xy, above
• But not xyz or xyz’ – they can
be expanded
18
Basic Concepts Underlying Automated Two-Level
Logic Minimization
• Definitions (cont)
– Essential prime implicant: The
only prime implicant that covers a
particular minterm in a function’s
on-set
G
yz
x
0
• Importance: We must include all
1
essential PIs in a function’s cover
x’y’
• In contrast, some, but not all, nonessential
essential PIs will be included
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Frank Vahid
not essential
y’z
00
01
11
10
1
1
0
0
0
1
1
1
xz
not essential
xy
essential
19
Automated Two-Level Logic Minimization Method
•
•
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Frank Vahid
Steps 1 and 2 are exact
Step 3: Hard. Checking all possibilities: exact, but computationally
expensive. Checking some but not all: heuristic.
20
Example of Automated Two-Level Minimization
I
• 1. Determine all
prime implicants
• 2. Add essential PIs
to cover
– Italicized 1s are thus
already covered
– Only one uncovered
1 remains
• 3. Cover remaining
minterms with nonessential PIs
– Pick among the two
possible PIs
Digital Design
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Frank Vahid
yz
00
01
x’z
11
10
0
1
1
1
0
1
1
0
0
1
x
(a)
y’z’ x’y’
I
yz
xz’
00
01
x’z
11
10
0
1
1
1
0
1
1
0
0
1
x
(b)
y’z’ x’y’
I
yz
xz’
00
01
x’z
11
10
0
1
1
1
0
1
1
0
0
1
x
(c)
y’z’
xz’
21
Problem with Methods that Enumerate all Minterms or
Compute all Prime Implicants
• Too many minterms for functions with many variables
– Function with 32 variables:
• 232 = 4 billion possible minterms.
• Too much compute time/memory
• Too many computations to generate all prime implicants
– Comparing every minterm with every other minterm, for 32
variables, is (4 billion)2 = 1 quadrillion computations
– Functions with many variables could requires days, months, years,
or more of computation – unreasonable
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22
Solution to Computation Problem
• Solution
– Don’t generate all minterms or prime implicants
– Instead, just take input equation, and try to “iteratively” improve it
– Ex: F = abcdefgh + abcdefgh’+ jklmnop
• Note: 15 variables, may have thousands of minterms
• But can minimize just by combining first two terms:
– F = abcdefg(h+h’) + jklmnop = abcdefg + jklmnop
Digital Design
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23
Two-Level Minimization using Iterative Method
• Method: Randomly apply “expand”
operations, see if helps
– Expand: remove a variable from a
term
I
yz
x
0
00
01
11
10
0
1
1
0
z
(a)
1
0
• Like expanding circle size on K-map
– e.g., Expanding x’z to z legal, but
expanding x’z to z’ not legal, in shown
function
– After expand, remove other terms
covered by newly expanded term
x’z
1
1
0
xy’z xyz
I
yz
x
0
00
01
11
10
0
1
1
0
x’
(b)
– Keep trying (iterate) until doesn’t help
x’z
1
0
1
xy’z
1
0
xyz
Ex:
F = abcdefgh + abcdefgh’+ jklmnop
F = abcdefg + abcdefgh’ + jklmnop
F = abcdefg + jklmnop
Digital Design
Copyright © 2006
Frank Vahid
24
Multi-Level Logic Optimization – Performance/Size
Tradeoffs
• We don’t always need the speed of two level logic
– Multiple levels may yield fewer gates
– Example
• F1 = ab + acd + ace  F2 = ab + ac(d + e) = a(b + c(d + e))
• General technique: Factor out literals – xy + xz = x(y+z)
b
4
c
a
c
e
4
b
a
d
a
4
c
6
6
F1
d
e
6
4
4
16 transistors
4 gate-delays
F1 = ab + acd + ace
F2 = a(b+c(d+e))
(a)
(b)
Digital Design
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Frank Vahid
F2
size
(transistors)
a
22 transistors
2 gate delays
F1
20
)15
t10rs
e
zsio
5
a
r(tnsis
1
F2
2
3
4
delay (gate-delays)
(c)
25
Multi-Level Example
• Q: Use multiple levels to reduce number of transistors for
– F1 = abcd + abcef
• A: abcd + abcef = abc(d + ef)
• Has fewer gate inputs, thus fewer transistors
a
a
b
c
d
a
b
c
e
f
18 transistors
3 gate delays
a
b
c
8
4
10
F1
6
d
e
4
4
4
f
F1 = abcd + abcef
(a)
Digital Design
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Frank Vahid
F2 = abc(d + ef)
(b)
size
(transistors)
22 transistors
2 gate delays
20
F1
F2
)15
F2
10
5
1 2 3 4
delay (gate-delays)
(c)
26
Multi-Level Example: Non-Critical Path
• Critical path: longest delay path to output
• Optimization: reduce size of logic on non-critical paths by using multiple
levels
Digital Design
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Frank Vahid
27
Automated Multi-Level Methods
• Main techniques use heuristic iterative methods
– Define various operations
• “Factor out”: xy + xz = x(y+z)
• Expand, and others
– Randomly apply, see if improves
• May even accept changes that worsen, in hopes eventually leads to
even better equation
• Keep trying until can’t find further improvement
– Not guaranteed to find best circuit, but rather a good one
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28
6.3
State Reduction (State Minimization)
• Goal: Reduce number of states in FSM without changing
behavior
– Fewer states potentially reduces size of state register
• Consider the two FSMs below with x=1, then 1, then 0, 0
Inputs: x; Outputs: y
x’
x’
x
x’
x
x
x’
x’
x
x
x
S0
S1
S2
S3
S0
S1
y=0
y=1
y=0
y=1
y=0
y=1
state
x
S0
S1
S1
x’
S2
S2
state
x
y
S0
S1
S1
a
S0
S0
y
For the same sequence of inputs,
the output of the two FSMs is the same
Digital Design
Copyright © 2006
Frank Vahid
x
y
if x = 1,1,0,0
then y = 0,1,1,0,0
29
State Reduction: Equivalent States
Two states are equivalent if:
1. They assign the same values to
outputs
– e.g. S0 and S2 both assign y to 0,
– S1 and S3 both assign y to 1
Inputs: x; Outputs: y
x’
x’
x
• starting from S1, y=1,1,0,0,…
• starting from S3, y=1,1,0,0,…
Digital Design
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Frank Vahid
x
x’
x
S0
S1
S2
S3
y=0
y=1
y=0
y=1
States S0 and S2 equivalent
States S1 and S3 equivalent
2. AND, for all possible sequences of
inputs, the FSM outputs will be the
same starting from either state
– e.g. say x=1,1,0,0,…
x’
x
a
x’
S0,
S2
y=0
x
x
x’
S1,
S3
y=1
30
State Reduction: Example with no Equivalencies
• Another example…
• State S0 is not equivalent with any
other state since its output (y=0)
differs from other states’ output
Inputs: x; Outputs: y
• Consider state S1 and S3
– Outputs are initially the same (y=1)
– From S1, when x=0, go to S2 where y=1
x’
x’
x’
x
x’
x
x
S0
S1
S2
S3
y=0
y=1
y=1
y=1
Start from S1, x=0
x’
x
x’
S0
y=0
S1
x’
x’
S1
y=1
S2
x
x
S3
y=1
y=1
x’
x
x’
x
x
a
x’
x
– From S3, when x=0, go to S0 where y=0
y=0
y=1
– Outputs differ, so S1 and S3 are not
Start from S3, x=0
equivalent.
S0
Digital Design
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Frank Vahid
x
x’
S2
y=1
x
S3
y=1
31
State Reduction with Implication Tables
• State reduction through visual inspection (what we did in
the last few slides) isn’t reliable and cannot be automated –
a more methodical approach is needed: implication tables
• Example:
S0
Inputs: x; Outputs: y
x’
x’
x
x’
S1
Redundant
S2
x’
x
x
x
S0
S1
S2
S3
y=0
y=1
y=0
y=1
Diagonal
S3
– To compare every pair of states, construct a
table of state pairs (above right)
– Remove redundant state pairs, and state pairs
along the diagonal since a state is equivalent
to itself (right)
S0 S1 S2 S3
S1
S2
S3
S0 S1 S2
Digital Design
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Frank Vahid
32
State Reduction with Implication Tables
• Mark (with an X) state pairs with different
outputs as non-equivalent:
– (S1,S0): At S1, y=1 and at S0, y=0. So S1
and S0 are non-equivalent.
– (S2, S0): At S2, y=0 and at S0, y=0. So we
don’t mark S2 and S0 now.
– (S2, S1): Non-equivalent
– (S3, S0): Non-equivalent
– (S3, S1): Don’t mark
– (S3, S2): Non-equivalent
• We can see that S2 & S0 might be
equivalent and S3 & S1 might be
equivalent, but only if their next states are
equivalent (remember the example from
two slides ago)
Digital Design
Copyright © 2006
Frank Vahid
Inputs: x; Outputs: y
x’
x’
x
x’
x
x
x’
x
S0
S1
S2
S3
y=0
y=1
y=0
y=1
S1
S2
a
S3
S0
S1
S2
33
State Reduction with Implication Tables
• We need to check each unmarked state
pair’s next states
• We can start by listing what each
unmarked state pair’s next states are for
every combination of inputs
– (S2, S0)
• From S2, when x=1 go to S3
From S0, when x=1 go to S1
So we add (S3, S1) as a next state pair
• From S2, when x=0 go to S2
From S0, when x=0 go to S0
So we add (S2, S0) as a next state pair
– (S3, S1)
Inputs: x; Outputs: y
x’
x’
x
x’
x
x
x’
x
S0
S1
S2
S3
y=0
y=1
y=0
y=1
S1
S2
(S3, S1)
(S2, S0)
a
(S3, S1)
(S0, S2)
S3
S0
S1
S2
• By a similar process, we add the next state
pairs (S3, S1) and (S0, S2)
Digital Design
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Frank Vahid
34
State Reduction with Implication Tables
• Next we check every unmarked
state pair’s next state pairs
• We mark the state pair if one of its
next state pairs is marked
– (S2, S0)
• Next state pair (S3, S1) is not marked
• Next state pair (S2, S0) is not marked
• So we do nothing and move on
– (S3, S1)
• Next state pair (S3, S1) is not marked
• Next state pair (S0, S2) is not marked
• So we do nothing and move on
Digital Design
Copyright © 2006
Frank Vahid
Inputs: x; Outputs: y
x’
x’
x
x’
x
x
x’
x
S0
S1
S2
S3
y=0
y=1
y=0
y=1
S1
S2
(S3, S1)
(S2, S0)
(S3, S1)
(S0, S2)
S3
S0
S1
S2
35
State Reduction with Implication Tables
• We just made a pass through the
implication table
– Make additional passes until no
change occurs
• Then merge the unmarked state
pairs – they are equivalent
Inputs: x; Outputs: y
x’
x
x
S1,S3
y=0
y=1
x
x’
x
S0
S1
S2
S3
y=0
y=1
y=0
y=1
S1
(S3, S1)
(S2, S0)
(S3, S1)
(S0, S2)
S3
S0
S0,S2
Digital Design
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Frank Vahid
x’
x’
x
S2
x’
x’
x
S1
S2
36
State Reduction with Implication Tables
Digital Design
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Frank Vahid
37
State Reduction Example
• Given FSM on the right
– Step 1: Mark state pairs having
different outputs as nonequivalent
Inputs: x; Outputs: y
x’
x’
x
x’
x
x
x’
x
S0
S1
S2
S3
y=0
y=1
y=1
y=1
S1
S2
a
S3
S0
Digital Design
Copyright © 2006
Frank Vahid
S1
S2
38
State Reduction Example
• Given FSM on the right
– Step 1: Mark state pairs having
different outputs as nonequivalent
– Step 2: For each unmarked state
pair, write the next state pairs for the
same input values
Inputs: x; Outputs: y
x’
x’
x
x
x’
x
S0
S1
S2
S3
y=0
y=1
y=1
y=1
S1
x=0
x=1
S2
(S2, S2)
(S3, S1)
S3
(S0, S2) (S0, S2)
(S3, S1) (S3, S3)
S0
Digital Design
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Frank Vahid
x’
x
S1
a
S2
39
Digital Design
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Frank Vahid
40
State Reduction Example
• Given FSM on the right
– Step 1: Mark state pairs having
different outputs as nonequivalent
– Step 2: For each unmarked state
pair, write the next state pairs for the
same input values
– Step 3: For each unmarked state
pair, mark state pairs having
nonequivalent next state pairs as
nonequivalent.
• Repeat this step until no change
occurs, or until all states are marked.
– Step 4: Merge remaining state pairs
Digital Design
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Frank Vahid
Inputs: x; Outputs: y
x’
x’
x
x’
x
x
x’
x
S0
S1
S2
S3
y=0
y=1
y=1
y=1
S1
S2
(S2, S2)
(S3, S1)
S3
(S0, S2) (S0, S2)
(S3, S1) (S3, S3)
S0
S1
a
S2
All state pairs are marked –
there are no equivalent
state pairs to merge
41
A Larger State Reduction Example
Inputs: x; Outputs: y
x’
S3
x’
S1
x’
y=0
x
x’
S0
y=0
S4
x
y=0
x’
x x
S2
S1
y=1
y=1
(S3,S4)
(S2,S1)
S2
S3 (S3,S2)
(S0,S1)
S4
– Step 1: Mark state pairs having different outputs as
nonequivalent
– Step 2: For each unmarked state pair, write the next state
pairs for the same input values
– Step 3: For each unmarked state pair, mark state pairs
having nonequivalent next state pairs as nonequivalent.
(S4,S3)
(S0,S0)
(S4,S2)
(S0,S1)
S0
S1
S2
S3
a
• Repeat this step until no change occurs, or until all states
are marked.
– Step 4: Merge remaining state pairs
Digital Design
Copyright © 2006
Frank Vahid
42
A Larger State Reduction Example
Inputs: x; Outputs: y
x’
S3
x’
S1
x’
y=0
x
x’
S0
y=0
S4
x
y=0
x’
x x
S2
S1
y=1
y=1
(S3,S4)
(S2,S1)
S2
S3 (S3,S2)
(S0,S1)
S4
– Step 1: Mark state pairs having different outputs as
nonequivalent
– Step 2: For each unmarked state pair, write the next state
pairs for the same input values
– Step 3: For each unmarked state pair, mark state pairs
having nonequivalent next state pairs as nonequivalent.
S0
x’
S2
S3
a
S3,S4
x
S0
y=0
Digital Design
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Frank Vahid
S1
Inputs: x; Outputs: y
• Repeat this step until no change occurs, or until all states
are marked.
– Step 4: Merge remaining state pairs
(S4,S3)
(S0,S0)
(S4,S2)
(S0,S1)
y=0
x’
x
x’
x
S1,S2
y=1
43
Need for Automation
• Automation needed
Inputs: x; Outputs: z
x
x
– Table for large FSM too big for
humans to work with
• n inputs: each state pair can have
next state pairs.
• 4 inputs  24=16 next state pairs
SB
z=0
z=1
x’
x
x
SC
x’
SD
x’ SG
z=0
x
z=1
z=0
x
x’
SE
z=0
x’
x’
SI
SH
SA
x’
2n
x
x’
x
SF
z=0
x
x’
x
x’
SJ
x
x’
x'
x’
z=1
z=0
x
x’
SL
x
z=0
SM
x’
z=0
SK
z=1
SO
z=1
SN
x
z=1
z=1
– 100 states would have table with 100*100=100,000 state pairs cells
– State reduction typically automated
• Often using heuristics to reduce compute time
Digital Design
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Frank Vahid
44
x
State Encoding
• Encoding: Assigning a unique
bit representation to each state
• Different encodings may
optimize size, or tradeoff size
and performance
• Consider 3-Cycle Laser Timer…
Inputs: b; Outputs: x
x=0
00
Off
b’
b
x=1
01 On1
x=1
11
10 On2
x=1
10
11 On3
– Example 3.7’s encoding: 15
gate inputs
– Try alternative encoding
•
•
•
•
x = s1 + s0
n1 = s0
n0 = s1’b + s1’s0
Only 8 gate inputs
Digital Design
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Frank Vahid
a
1
1
1
1
0
0
0
0
45
State Encoding: One-Hot Encoding
•
Inputs: none; Outputs: x
x=0
One-hot encoding
– One bit per state – a bit being ‘1’
corresponds to a particular state
– Alternative to minimum bit-width
encoding in previous example
– For A, B, C, D: A: 0001, B: 0010, C:
0100, D: 1000
•
x=1
A 00
0001
D 11
1000
B 01
0010
x=1
C 10
0100
x=1
a
Example: FSM that outputs 0, 1, 1, 1
– Equations if one-hot encoding:
• n3 = s2; n2 = s1; n1 = s0; x = s3 +
s2 + s1
– Fewer gates and only one level of
logic – less delay than two levels, so
faster clock frequency
x
x
n1
8
binary
6
4 one-hot
2
Digital Design
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Frank Vahid
1 2 3 4
delay (gate-delays)
s3
s1
s0
clk
clk
s2
s1
s0
n0
State register
State register
n0
n1
n2
n3
46
One-Hot Encoding Example:
Three-Cycles-High Laser Timer
• Four states – Use four-bit one-hot
encoding
– State table leads to equations:
•
•
•
•
•
x = s3 + s2 + s1
n3 = s2
n2 = s1
n1 = s0*b
n0 = s0*b’ + s3
Inputs: b; Outputs: x
x=0
0001
Off
b’
b
x=1
x=1
x=1
0010
On1
0100
On2
1000
On3
a
– Smaller
• 3+0+0+2+(2+2) = 9 gate inputs
• Earlier binary encoding (Ch 3):
15 gate inputs
– Faster
• Critical path: n0 = s0*b’ + s3
• Previously: n0 = s1’s0’b + s1s0’
• 2-input AND slightly faster than 3input AND
Digital Design
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Frank Vahid
47
Output Encoding
• Output encoding: Encoding
method where the state
encoding is same as the
output values
– Possible if enough outputs, all
states with unique output values
Use the output values
as the state encoding
Inputs: none; Outputs: x,y
xy=00
A 00
D 11
B 01
C 10
xy=11
Digital Design
Copyright © 2006
Frank Vahid
xy=01
a
xy=10
48
Output Encoding Example: Sequence Generator
w
x
y
z
Inputs: none; Outputs: w, x, y, z
wxyz=0001
wxyz=1000
A
D
B
C
wxyz=0011
wxyz=1100
• Generate sequence 0001, 0011, 1110,
1000, repeat
– FSM shown
• Use output values as state encoding
• Create state table
• Derive equations for next state
s3
clk
s2
s1 s0
State register
n3
n2
n1
n0
– n3 = s1 + s2; n2 = s1; n1 = s1’s0; n0 = s1’s0
+ s3s2’
Digital Design
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Frank Vahid
49
Moore vs. Mealy FSMs
Combinational
logic
I
O
S
clk
State register
N
Moore
Mealy
a
• FSM implementation architecture
– State register and logic
– More detailed view
• Next state logic – function of present state and FSM
inputs
• Output logic
– If function of present state only – Moore FSM
– If function of present state and FSM inputs – Mealy FSM
Digital Design
Copyright © 2006
Frank Vahid
Inputs: b; Outputs: x
/x=0
S0
S1
b/x=1
b’/x=0
Graphically: show outputs with
arcs, not with states
50
Mealy FSMs May Have Fewer States
Inputs: enough (bit)
Outputs: d, clear (bit)
Inputs: enough (bit)
Outputs: d, clear (bit)
/d=0, clear=1
Moore
Init
d=0
clear=1
Wait
Init
Mealy
Wait
enough’
enough’
enough
Disp
enough/d=1
d=1
clk
clk
Inputs: enough
Inputs: enough
State:
I W W D I
State:
Outputs: clear
Outputs: clear
d
d
(a)
I W W I
(b)
• Soda dispenser example: Initialize, wait until enough, dispense
– Moore: 3 states; Mealy: 2 states
Digital Design
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Frank Vahid
51
Mealy vs. Moore
• Q: Which is Moore,
and which is Mealy?
• A: Mealy on left,
Moore on right
– Mealy outputs on
arcs, meaning
outputs are function
of state AND INPUTS
– Moore outputs in
states, meaning
outputs are function
of state only
Digital Design
Copyright © 2006
Frank Vahid
Inputs: b; Outputs: s1, s0, p
Time
b’/s1s0=00, p=0
b/s1s0=00, p=1
Alarm
b’/s1s0=01, p=0
b/s1s0=01, p=1
Date
b’/s1s0=10, p=0
b/s1s0=10, p=1
Stpwch
b’/s1s0=11, p=0
b/s1s0=11, p=1
Mealy
Inputs: b; Outputs: s1, s0, p
Time
b’
s1s0=00, p=0
b
S2 s1s0=00, p=1
b
b’
Alarm
s1s0=01, p=0
b
S4 s1s0=01, p=1
b
Date
b’
s1s0=10, p=0
b
S6 s1s0=10, p=1
b
b’
Stpwch
s1s0=11, p=0
b
S8 s1s0=11, p=1
Moore
52
Mealy vs. Moore Example: Beeping Wristwatch
• Button b
– Sequences mux select lines
s1s0 through 00, 01, 10, and
11
• Each value displays different
internal register
– Each unique button press
should cause 1-cycle beep,
with p=1 being beep
• Must wait for button to be
released (b’) and pushed
again (b) before sequencing
• Note that Moore requires
unique state to pulse p, while
Mealy pulses p on arc
• Tradeoff: Mealy’s pulse on p
may not last one full cycle
Digital Design
Copyright © 2006
Frank Vahid
Inputs: b; Outputs: s1, s0, p
Time
b’/s1s0=00, p=0
b/s1s0=00, p=1
Alarm
b’/s1s0=01, p=0
b/s1s0=01, p=1
Date
b’/s1s0=10, p=0
b/s1s0=10, p=1
Stpwch
b’/s1s0=11, p=0
b/s1s0=11, p=1
Mealy
Inputs: b; Outputs: s1, s0, p
Time
b’
s1s0=00, p=0
b
S2 s1s0=00, p=1
b
b’
Alarm
s1s0=01, p=0
b
S4 s1s0=01, p=1
b
Date
b’
s1s0=10, p=0
b
S6 s1s0=10, p=1
b
b’
Stpwch
s1s0=11, p=0
b
S8 s1s0=11, p=1
Moore
53
Mealy vs. Moore Tradeoff
• Mealy outputs change mid-cycle if input changes
– Note earlier soda dispenser example
• Mealy had fewer states, but output d not 1 for full cycle
– Represents a type of tradeoff
Inputs: enough (bit)
Outputs: d, clear (bit)
Inputs: enough (bit)
Outputs: d, clear (bit)
/d=0, clear=1
Moore
Init
d=0
clear=1
Wait
Init
Mealy
Wait
enough’
enough’
enough
Disp
enough/d=1
d=1
clk
clk
Inputs: enough
Inputs: enough
State:
I W W D I
State:
Outputs: clear
Outputs: clear
d
d
Digital Design
Copyright © 2006
Frank Vahid
(a)
I W W I
54
(b)
Implementing a Mealy FSM
• Straightforward
– Convert to state table
– Derive equations for each
output
– Key difference from
Moore: External outputs
(d, clear) may have
different value in same
state, depending on input
values
Digital Design
Copyright © 2006
Frank Vahid
Inputs: enough (bit)
Outputs: d, clear (bit)
/ d=0, clear=1
Init
Wait
enough’/d=0
enough/d=1
55
Mealy and Moore can be Combined
• Final note on Mealy/Moore
– May be combined in same FSM
Inputs: b; Outputs: s1, s0, p
Time
b’/p=0
s1s0=00
b/p=1
b’/p=0
Alarm
s1s0=01
b/p=1
Combined
b’/p=0
Moore/Mealy
Date
FSM for beeping
s1s0=10
b/p=1
wristwatch
example
b’/p=0
Stpwch
s1s0=11
b/p=1
Digital Design
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Frank Vahid
56
Datapath Component Tradeoffs
6.4
• Can make some components faster (but bigger), or smaller (but
slower), than the straightforward components we built in Ch 4
• We’ll build
– A faster (but bigger) adder than the carry-ripple adder
– A smaller (but slower) multiplier than the array-based multiplier
• Could also do for the other Ch 4 components
Digital Design
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Frank Vahid
57
Faster Adder
• Built carry-ripple adder in Ch 4
a3 b3
– Similar to adding by hand, column by column
– Con: Slow
• Output is not correct until the carries have
rippled to the left
• 4-bit carry-ripple adder has 4*2 = 8 gate delays
a2 b2
a1 b1
a0 b0
cin
4-bit adder
cout
s3
s2
s1
s0
carries:
c3
c2
c1
cin
B:
b3
b2
b1
b0
A:
+ a3
a2
a1
a0
s2
s1
s0
– Pro: Small
cout
s3
a
• 4-bit carry-ripple adder has just 4*5 = 20 gates
a3 b3
a2 b2
a1b1
a0 b0
ci
a
FA
FA
co
Digital Design
Copyright © 2006
Frank Vahid
s3
FA
s2
FA
s1
s0
58
Faster Adder
a3 b3 a2 b2 a1b1 a0 b0
ci
• Faster adder – Use two-level
combinational logic design process
– Recall that 4-bit two-level adder was big
– Pro: Fast
Two-level: AND level
followed by ORs
• 2 gate delays
– Con: Large
• Truth table would have 2(4+4) =256 rows
• Plot shows 4-bit adder would use about
500 gates
• Is there a compromise design?
– Between 2 and 8 gate delays
– Between 20 and 500 gates
Digital Design
Copyright © 2006
Frank Vahid
co s3
s2 s1 s0
2
4
10000
8000
6000
rs
to4000
nsis
ra
T2000
0
1
3
N
5
6
7
59
8
Faster Adder – (Bad) Attempt at “Lookahead”
• Idea
– Modify carry-ripple adder – For a stage’s carry-in, don’t wait for carry
to ripple, but rather directly compute from inputs of earlier stages
• Called “lookahead” because current stage “looks ahead” at previous
stages rather than waiting for carry to ripple to current stage
a3a3
b3 b3
a2
a2
b2b2
a1
a1b1
b1
a0 b0
a0c0
b0
ci
a
look
ahead
look
ahead
c3
FA
FA
c4 co
cout
Digital Design
Copyright © 2006
Frank Vahid
c2
FA
s3 3
stage
s3
look
ahead
c1
FA
stage 2s2
s2
FA c0
stage 1 s1
s1
stage 0
s0
s0
Notice – no rippling of carry
60
Faster Adder – (Bad) Attempt at “Lookahead”
•
Want each stage’s carry-in bit to be function of external inputs only (a’s, b’s, or c0)
c3
a3b3
a2b2
c2
a1b1
c1
a0 b0
c0
a
FA
FA
co
•
co2
s3
s1
FA
co0
s0
Recall full-adder equations:
– s = a xor b
– c = bc + ac + ab
a3b3
a2b2
a1b1
look
ahead
look
ahead
c3
s3
c1
stage 2
s2
Stage 1: c1=co0
look
ahead
c2
stage 3
Stage 0: Carry-in is already an
external input: c0
a0b0 c0
stage 1
s1
Digital Design
Copyright © 2006
Frank Vahid
Stage 2: c2=co1
c0
FA
c4
cout
s2
FA
co1
co0= b0c0 + a0c0 + a0b0
c1 = b0c0 + a0c0 + a0b0
co1 = b1c1 + a1c1 + a1b1
c2 = b1c1 + a1c1 + a1b1
c2 = b1(b0c0 + a0c0 + a0b0) + a1(b0c0 + a0c0 + a0b0) +a1b1
c2 = b1b0c0 + b1a0c0 + b1a0b0 + a1b0c0 + a1a0c0 + a1a0b0 + a1b1
stage 0
s0
Continue for c3
61
Faster Adder – (Bad) Attempt at “Lookahead”
a3b3
a2b2
a1b1
a0b0 c0
• Carry lookahead logic
function of external inputs
– No waiting for ripple
look
ahead
look
ahead
• Problem
– Equations get too big
– Not efficient
– Need a better form of
lookahead
c3
look
ahead
c2
c1
c0
FA
stage 3
c4
cout
s3
stage 2
s2
stage 1
s1
stage 0
s0
c1 = b0c0 + a0c0 + a0b0
c2 = b1b0c0 + b1a0c0 + b1a0b0 + a1b0c0 + a1a0c0 + a1a0b0 + a1b1
c3 = b2b1b0c0 + b2b1a0c0 + b2b1a0b0 + b2a1b0c0 + b2a1a0c0 + b2a1a0b0 + b2a1b1 +
a2b1b0c0 + a2b1a0c0 + a2b1a0b0 + a2a1b0c0 + a2a1a0c0 + a2a1a0b0 + a2a1b1 + a2b2
Digital Design
Copyright © 2006
Frank Vahid
62
Better Form of Lookahead
• Have each stage compute two terms
– Propagate: P = a xor b
– Generate: G = ab
• Compute lookahead from P and G terms, not from external inputs
– Why P & G? Because the logic comes out much simpler
• Very clever finding; not particularly obvious though
• Why those names?
– G: If a and b are 1, carry-out will be 1 – “generate” a carry-out of 1 in this case
– P: If only one of a or b is 1, then carry-out will equal the carry-in – propagate the
carry-in to the carry-out in this case
cin
carries: c4 c3 c2 c1 c0
c1
B:
b3 b2 b1 b0
A:
+ a3 a2 a1 a0
s3 s2 s1 s0
cout
0
1
+ 1
0
(a)
Digital Design
Copyright © 2006
Frank Vahid
1
1
1
1
+ 1
1
1
1
0
+ 1
0
1
1
1
+ 0
0
c0
b0
a0
if a0b0 = 1
if a0xor b0 = 1
then c1 = 1
then c1 = 1 if c0 = 1
(call this G:Generate) (call this P: Propagate)
63
Better Form of Lookahead
a3 b3
a3 b3
a2 b2
Half-adder
a2 b2
a1 b1 a1 b1
“Bad” lookahead
Half-adder
Half-adder
look
ahead
look
ahead
a
c3
G3
P3FA
c3
G2
Carry-loo kahead lo gic
c4
stage 3
cout
cout
s3
c2
c2
P2
s3
• With P & G, the carry lookahead
equations are much simpler
– Equations before plugging in
•
•
•
•
c1 = G0 + P0c0
c2 = G1 + P1c1
c3 = G2 + P2c2
cout = G3 + P3c3
Digital Design
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Frank Vahid
look
ahead
c1
G1
(b)
cin
Half-adder
P1
c1
stage 2
s2
s2
a0 b0 c0
a0 b0
G0
c0
P0
stage 1
s1
s1
c0
stage 0
s0
s0
After plugging in:
c1 = G0 + P0c0
a
c2 = G1 + P1c1 = G1 + P1(G0 + P0c0)
c2 = G1 + P1G0 + P1P0c0
c3 = G2 + P2c2 = G2 + P2(G1 + P1G0 + P1P0c0)
c3 = G2 + P2G1 + P2P1G0 + P2P1P0c0
cout = G3 + P3G2 + P3P2G1 + P3P2P1G0 +
P3P2P1P0c0
Much simpler than the “bad” lookahead
64
Call this
sum/propagate/generate
(SPG) block
Better Form of Lookahead
a
a3 b3
Half-adder
a2 b2
SPG
block
G3
P3
c3
Carry-loo kahead lo gic
cout
a1 b1
Half-adder
G2
Half-adder
P2
s3
c2
s2
P3 G3
a0 b0
G1
P2 G2
Half-adder
P1
(b)
cin
c1
G0
P0
s0
s1
P1 G1
c0
P0
G0
c0
Carry-loo kahead lo gic
a
Stage 4
Stage 3
Stage 2
Stage 1
c1 = G0 + P0c0
c2 = G1 + P1G0 + P1P0c0
c3 = G2 + P2G1 + P2P1G0 + P2P1P0c0
cout = G3 + P3G2 + P3P2G1 + P3P2P1G0 + P3P2P1P0c0
(c)
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Frank Vahid
65
Carry-Lookahead Adder -- High-Level View
a3
a2
b3
a
b
cin
SPG block
P
G
P3
G3
a
b
cin
SPG block
P
c3 P2
cout
cout
a1
b2
s3
G
a
P
b
cin
SPG block
G
s2
– Each stage has SPG block with 2 gate levels
– Carry-lookahead logic quickly computes the
carry from the propagate and generate bits
using 2 gate levels inside
• Reasonable number of gates -- 4-bit adder
has only 26 gates
a
P
G2
c2 P1
G1
4-bit carry-lookahead logic
• Fast -- only 4 gate delays
Digital Design
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Frank Vahid
a0
b1
c1 P0
s1
b0
c0
b
cin
SPG block
G
G0
s0
• 4-bit adder comparison
(gate delays, gates)
– Carry-ripple: (8, 20)
– Two-level: (2, 500)
– CLA: (4, 26)
o Nice compromise
66
Carry-Lookahead Adder – 32-bit?
• Problem: Gates get bigger in each stage
– 4th stage has 5-input gates
– 32nd stage would have 33-input gates
• Too many inputs for one gate
• Would require building from smaller gates,
meaning more levels (slower), more gates
(bigger)
• One solution: Connect 4-bit CLA adders in
ripple manner
– But slow (4 + 4 + 4 + 4 gate delays)
a15-a12
b15-b12
a3a2a1a0 b3b2b1b0
4-bit adder cin
cout s3s2s1s0
cout
s15-s12
Digital Design
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Frank Vahid
a11-a8
b11-b8
a3a2a1a0 b3b2b1b0
4-bit adder cin
cout s3s2s1s0
s11-s8
a7a6a5a4 b7b6b5b4
a3a2a1a0 b3b2b1b0
a3a2a1a0 b3b2b1b0
4-bit adder cin
a3a2a1a0 b3b2b1b0
4-bit adder cin
cout s3s2s1s0
cout s3s2s1s0
s7s6s5s4
s3s2s1s0
67
Hierarchical Carry-Lookahead Adders
• Better solution -- Rather than rippling the carries, just repeat the carrylookahead concept
– Requires minor modification of 4-bit CLA adder to output P and G
These use carry-lookahead internally
a15-a12
b15-b12
a11-a8
b11-b8
a7a6a5a4
b7b6b5b4
a3a2a1a0
b3b2b1b0
a3a2a1a0
b3b2b1b0
a3a2a1a0
b3b2b1b0
a3a2a1a0
b3b2b1b0
a3a2a1a0
b3b2b1b0
4-bit adder
P G cout
cin
4-bit adder
P G cout s3s2 s1s0
s3s2 s1s0
P3G3
cin
c3 P2 G2
P G cout
s15-s12
4-bit adder
cin
4-bit adder
P G cout s3s2 s1s0
c2 P1 G1
4-bit carry-lookahead logic
s11-s18
P G cout s3s2 s1s0
c1 P0 G0
s7-s4
s3-s0
Second level of carry-lookahead
P2 G2
P3 G3
Carry lookahead logic
P1 G1
P0 G0
c0
Same lookahead logic as
inside the 4-bit adders
a
Digital Design
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Frank Vahid
cin
Stage 4
cout
Stage 3
c3
Stage 2
c2
Stage 1
c1
68
Hierarchial Carry-Lookahead Adders
•
•
Hierarchical CLA concept can be applied for larger adders
32-bit hierarchical CLA
– Only about 8 gate delays (2 for SPG block, then 2 per CLA level)
– Only about 14 gates in each 4-bit CLA logic block
a
Q: How many gate
delays for 64-bit
hierarchical CLA,
using 4-bit CLA logic?
A: 16 CLA-logic blocks
in 1st level, 4 in 2nd, 1
in 3rd -- so still just 8
gate delays (2 for
SPG, and 2+2+2 for
CLA logic). CLA is a
very efficient method.
Digital Design
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Frank Vahid
69
Carry Select Adder
• Another way to compose adders
– High-order stage -- Compute result for carry in of 1 and of 0
• Select based on carry-out of low-order stage
• Faster than pure rippling
Operate in parallel
suppose =1
Digital Design
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Frank Vahid
70
Adder Tradeoffs
size
carry-lookahead
multilevel
carry-lookahead
carry-select
carryripple
delay
• Designer picks the adder that satisfies particular delay and
size requirements
– May use different adder types in different parts of same design
• Faster adders on critical path, smaller adders on non-critical path
Digital Design
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Frank Vahid
71
Smaller Multiplier
• Multiplier in Ch 4 was array style
– Fast, reasonable size for 4-bit: 4*4 = 16 partial product AND terms, 3 adders
– Rather big for 32-bit: 32*32 = 1024 AND terms, and 31 adders
a
32-bit adder would have 1024 gates here ...
a3
a2
a0
pp1
b0
a1
pp2
b1
pp3
b2
pp4
b3
0
0
+ (5-bit)
00
a
+ (6-bit)
0 00
... and 31 adders
here (big ones, too)
+ (7-bit)
Digital Design
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Frank Vahid
p7..p0
72
Smaller Multiplier -- Sequential (Add-and-Shift) Style
• Smaller multiplier: Basic idea
– Don’t compute all partial products simultaneously
– Rather, compute one at a time (similar to by hand), maintain running
sum
(running sum)
Step 1
0110
+ 0 0 11
Step 2
0110
+ 0 01 1
Step 3
0110
+ 00 1 1
Step 4
0110
+ 00 11
0000
00110
010010
0010010
+ 0000
0010010
+ 0000
00010010
(partial product) + 0 1 1 0
(new running sum) 0 0 1 1 0
Digital Design
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Frank Vahid
+0 1 1 0
010010
a
73
Smaller Multiplier -- Sequential (Add-and-Shift) Style
multiplier
• Design circuit that
computes one partial
product at a time, adds to
running sum
– Note that shifting
running sum right
(relative to partial
product) after each step
ensures partial product
added to correct running
sum bits
Step 1
0110
+ 0011
0000
+ 0110
00110
Step 2
0110
+ 0 01 1
00110
+0 1 1 0
010010
Digital Design
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Frank Vahid
Step 3
0110
+ 001 1
010010
+ 0000
0010010
multiplicand
multiplicand
register (4)
load
mdld
mrld
o
rtnller
mr3
o
c mr2
mr1
4-bit adder
multiplier
register (4)
load
mr0
rsload
rsclear
rsshr
start
Step 4
0110
+ 0011
(running sum)
0010010
(partial product)
+ 0000
0 0 0 1 0 0 1 0 (new running sum)
load
clear
shr
running sum
register (8)
a
product
74
Smaller Multiplier -- Sequential Style: Controller
controller
start’
mdld
start
mdld = 1
mrld = 1
rsclear = 1
rsshr=1
mr0’
mr1’
mrld
rsshr=1
mr2’
rsshr=1
mr3
mr2
mr1
mr0
rsload
rsclear
rsshr
rsshr=1
mr3’
a
mr0
mr1
rsload=1
mr2
rsload=1
mr3
rsload=1
rsload=1
start
Wait for start=1
Looks at multiplier one bit at a
time
– Adds partial product
(multiplicand) to running sum if
present multiplier bit is 1
– Then shifts running sum right
one position
multiplicand
multiplicand
register (4)
load
controller
•
•
multiplier
Vs. array-style:
a
Pro: small
• Just three registers,
adder, and controller
Con: slow
• 2 cycles per multiplier
bit
• 32-bit: 32*2=64 cycles
(plus 1 for init.)
mdld
mrld
mr3
mr2
mr1
mr0
rsload
rsclear
rsshr
0110
4-bit adder
multiplier
register (4)
load
0011
a
load
clear
shr
running sum
register (8)
10010
00110
000
0110
00000000
0000
0010010
00010010
010010
000
start
Digital Design
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Frank Vahid
Correct product
product
75
RTL Design Optimizations and Tradeoffs
6.5
• While creating datapath during RTL design, there are
several optimizations and tradeoffs, involving
–
–
–
–
–
–
Pipelining
Concurrency
Component allocation
Operator binding
Operator scheduling
Moore vs. Mealy high-level state machines
Digital Design
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Frank Vahid
76
Pipelining
• Intuitive example: Washing dishes
with a friend, you wash, friend dries
– You wash plate 1
– Then friend dries plate 1, while you wash
plate 2
– Then friend dries plate 2, while you wash
plate 3; and so on
– You don’t sit and watch friend dry; you
start on the next plate
Time
Without pipelining:
W1 D1 W2 D2 W3 D3
a
With pipelining:
W1 W2 W3
D1 D2 D3
“Stage 1”
“Stage 2”
• Pipelining: Break task into stages,
each stage outputs data for next
stage, all stages operate concurrently
(if they have data)
Digital Design
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Frank Vahid
77
Pipelining Example
X
Y
clk
+
2ns
2ns
+
W
Z
Stage 1
W
X
Y
+
Z
Longest path
is only 2 ns
+
2ns 1 2ns
stage
2nsLongest path
clk
pipeline
registers
a
+
2ns
S
So minimum clock
period is 4ns
clk
S
S(0)
Stage 2
is 2+2 = 4 ns
+
2ns
stage 2S
So minimum clock
period is 2ns
clk
S(1)
S
S(0) S(1)
• S = W+X+Y+Z
• Datapath on left has critical path of 4 ns, so fastest clock period is 4 ns
– Can read new data, add, and write result to S, every 4 ns
• Datapath on right has critical path of only 2 ns
– So can read new data every 2 ns – doubled performance (sort of...)
Digital Design
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78
Pipelining Example
• Pipelining requires refined definition of performance
– Latency: Time for new data to result in new output data (seconds)
– Throughput: Rate at which new data can be input (items / second)
– So pipelining above system
• Doubled the throughput, from 1 item / 4 ns, to 1 item / 2 ns
• Latency stayed the same: 4 ns
Digital Design
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Frank Vahid
79
Pipeline Example: FIR Datapath
• 100-tap FIR filter: Row of
100 concurrent multipliers,
followed by tree of adders
– Assume 20 ns per multiplier
– 14 ns for entire adder tree
– Critical path of 20+14 = 34 ns
• Add pipeline registers
– Longest path now only 20 ns
– Clock frequency can be nearly
doubled
• Great speedup with minimal
extra hardware
Digital Design
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Frank Vahid
80
Concurrency
• Concurrency: Divide task into
Task
subparts, execute subparts
simultaneously
– Dishwashing example: Divide stack
into 3 substacks, give substacks to
3 neighbors, who work
simultaneously -- 3 times speedup
(ignoring time to move dishes to
neighbors' homes)
– Concurrency does things side-byside; pipelining instead uses stages
(like a factory line)
– Already used concurrency in FIR
filter -- concurrent multiplications
Digital Design
Copyright © 2006
Frank Vahid
*
a
Concurrency
Can do both, too
Pipelining
*
*
81
Concurrency Example: SAD Design Revisited
• Sum-of-absolute differences video compression example (Ch 5)
– Compute sum of absolute differences (SAD) of 256 pairs of pixels
– Original : Main loop did 1 sum per iteration, 256 iterations, 2 cycles per iter.
AB_addr A_data B_data
!go
S0
go
S1
(i<256)’
sum = 0
i=0
i_lt_256
<256
i_inc
i_clr
8
9
8
–
i
8
S2
sum_ld
i<256
sum=sum+abs(A[i]-B[i])
S3
i=i+1
sum_clr
sum
abs
8
32 32
sad_reg_ld
sad_reg
S4
32
sad_reg=sum
Datapath
+
32
sad
256 iters.*2 cycles/iter. = 512 cycles
Digital Design
Copyright © 2006
Frank Vahid
-/abs/+ done in 1 cycle,
but done 256 times
82
Concurrency Example: SAD Design Revisited
• More concurrent design
– Compute SAD for 16 pairs concurrently, do 16 times to compute all
16*16=256 SADs.
– Main loop does 16 sums per iteration, only 16 iters., still 2 cycles per iter.
AB_addr
go AB_rd
i_lt_16
S1
i_lt_16’
New: 16*2 = 32 cycles
Orig: 256*2 = 512 cycles
S0
go
!go
sum_clr=1
i_clr=1
S2
i_lt_16
AB_rd=1
sum_ld=1
i_inc=1
!(i_lt_16)
S4
A0 B0 A1 B1 A14 B14 A15 B15
a
<16
i_inc
i_clr
i
–
–
–
–
sum
abs
abs
abs
abs
sum_ld
sum_clr
sad_reg_ld
+
Adder tree to
sum 16 values
+
Controller
Digital Design
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Frank Vahid
16 absolute
values
+
sad_reg
sad_reg_ld=1
16 subtractors
+
Datapath
a
sad
All -/abs/+’s shown done in 1
cycle, but done only 16 times
83
Concurrency Example: SAD Design Revisited
• Comparing the two designs
– Original: 256 iterations * 2 cycles/iter = 512 cycles
– More concurrent: 16 iterations * 2 cycles/iter = 32 cycles
– Speedup: 512/32 = 16x speedup
• Versus software
– Recall: Estimated about 6 microprocessor cycles per iteration
• 256 iterations * 6 cycles per iteration = 1536 cycles
• Original design speedup vs. software: 1536 / 512 = 3x
– (assuming cycle lengths are equal)
!(i_lt_16)design’s speedup vs. software: 1536 / 32 = 48x
• Concurrent
– 48x is very significant – quality of video may be much better
Digital Design
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Frank Vahid
84
Component Allocation
• Another RTL tradeoff: Component allocation – Choosing a particular
set of functional units to implement a set of operations
– e.g., given two states, each with multiplication
• Can use 2 multipliers (*)
• OR, can instead use 1 multiplier, and 2 muxes
• Smaller size, but slightly longer delay due to the mux delay
A
B
t1 = t2*t3
t4 = t5*t6
FSM-A: (t1ld=1) B: (t4ld=1)
t2
t3
t5
t6
A: (sl=0; sr=0; t1ld=1)
B: (sl=1; sr=1; t4ld=1)
t2 t5 t3 t6
sl 2×1 2×1 sr
a
2 mul
1 mul
*
*
t1
Digital Design
Copyright © 2006
Frank Vahid
*
t4
(a)
t1
t4
delay
(c)
(b)
85
Operator Binding
• Another RTL tradeoff: Operator binding – Mapping a set of operations
to a particular component allocation
A
B
C
A
B
C
t1 = t2* t3
t4 = t5* t6
t7 = t8* t3
t1 = t2* t3
t4 = t5* t6
t7 = t8* t3
t2
t3
t5 t8
sl 2x1
2 multipliers
MULA
allocated
t1
2 muxes
vs.
2x1 sr
1 mux
t2 t8
t4
t7
t3
t5
t6
e
zsi
sl 2x1
MULB
Binding 1
Digital Design
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Frank Vahid
t6 t3
MULA
t1
size
– Note: operator/operation mean behavior (multiplication, addition), while
component (aka functional unit) means hardware (multiplier, adder)
– Different bindings may yield different size or delay
MULB
t7
Binding 1
Binding 2
a
delay
t4
Binding 2
86
Operator Scheduling
• Yet another RTL tradeoff: Operator scheduling –
Introducing or merging states, and assigning operations to
those states.
A
A
C
B
(some
t1 = t2*t3
operations) t4 = t5*t6
B2
B
(some
t1 = t2* t3
operations) t4 = t5* t6
(some
operations)
t4 = t5* t6
t2 t5
t3
t6
sl 2x1
*
t1
Digital Design
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Frank Vahid
3-state schedule
t4
si
e
z
size
*
t5
smaller
(only 1 *)
4-state schedule
(some
operations)
t3 t6
a
t2
C
2x1 sr
but more
delay due to
muxes
*
t1
t4
delay
87
Operator Scheduling Example: Smaller FIR Filter
• 3-tap FIR filter design in Ch 5: Only one state – datapath computes new
Y every cycle
– Used 3 multipliers and 2 adders; can we reduce the design’s size?
y(t) = c0*x(t) + c1*x(t-1) + c2*x(t-2)
Inputs: X (N bits)
Outputs: Y (N bits)
Local registers:
xt0, xt1, xt2 (N bits)
CL
3-tap FIR filter
e
Ca1
Ca0
3
2x4 2
1
0
C
S1
xt0 = X
xt1 = xt0
xt2 = xt1
Y = xt0*c0
+ xt1*c1
+ xt2*c2
x(t)
c0
xt0
x(t-1)
c1
xt1
x(t-2)
c2
xt2
X
clk
*
*
+
*
+
yreg
Y
Digital Design
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Frank Vahid
88
Operator Scheduling Example: Smaller FIR Filter
• Reduce the design’s size by re-scheduling the operations
– Do only one multiplication operation per state
Inputs: X (N bits)
Outputs: Y (N bits)
Local registers:
xt0, xt1, xt2 (N bits)
xt0 = X
xt1 = xt0
xt2 = xt1
Y = xt0*c0
+ xt1*c1
+ xt2*c2
S1
(a)
Inputs: X (N bits)
Outputs: Y (N bits)
Local registers:
xt0, xt1, xt2, sum (N bits)
sum = 0
xt0 = N
S1
xt1 = xt0
xt2 = xt1
S2
sum = sum + xt0 * c0
a
S3
sum = sum +xt1 * c1
S4
sum = sum + xt2 * c2
S5
Y = sum
y(t) = c0*x(t) + c1*x(t-1) + c2*x(t-2)
Digital Design
Copyright © 2006
Frank Vahid
(b)
89
Operator Scheduling Example: Smaller FIR Filter
• Reduce the design’s size by re-scheduling the operations
– Do only one multiplication (*) operation per state, along with sum (+)
Inputs: X (N bits)
Outputs: Y (N bits)
Local registers:
xt0, xt1, xt2, sum (N bits)
sum = 0
xt0 = X
S1
xt1 = xt0
xt2 = xt1
X
xt0
c0
c1
xt1
x_ld
sum = sum + xt0 * c0
yreg
3x1
3x1
mul_s0
S3
*
y_ld
S4
S5
Y = sum
Digital Design
Copyright © 2006
Frank Vahid
Y
a
sum = sum + xt1 * c1
sum = sum + xt2 * c2
c2
clk
mul_s1
S2
xt2
MAC
+
sum
Multiplyaccumulate: a
common datapath
component
90
Operator Scheduling Example: Smaller FIR Filter
• Many other options exist
between fully-concurrent and
fully-serialized
Digital Design
Copyright © 2006
Frank Vahid
concurrent FIR
compromises
size
– e.g., for 3-tap FIR, can use 1, 2,
or 3 multipliers
– Can also choose fast array-style
multipliers (which are concurrent
internally) or slower shift-andadd multipliers (which are
serialized internally)
– Each options represents
compromises
serial
FIR
delay
91
More on Optimizations and Tradeoffs
6.6
• Serial vs. concurrent computation has been a common tradeoff
theme at all levels of design
– Serial: Perform tasks one at a time
– Concurrent: Perform multiple tasks simultaneously
• Combinational logic tradeoffs
– Concurrent: Two-level logic (fast but big)
– Serial: Multi-level logic (smaller but slower)
• abc + abd + ef  (ab)(c+d) + ef – essentially computes ab first (serialized)
• Datapath component tradeoffs
– Serial: Carry-ripple adder (small but slow)
– Concurrent: Carry-lookahead adder (faster but bigger)
• Computes the carry-in bits concurrently
– Also multiplier: concurrent (array-style) vs. serial (shift-and-add)
• RTL design tradeoffs
– Concurrent: Schedule multiple operations in one state
– Serial: Schedule one operation per state
Digital Design
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Frank Vahid
92
Higher vs. Lower Levels of Design
• Optimizations and tradeoffs at higher levels typically have
greater impact than those at lower levels
– RTL decisions impact size/delay more than gate-level decisions
Spotlight analogy: The lower you
are, the less solution landscape is
illuminated (meaning possible)
Digital Design
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Frank Vahid
93
Algorithm Selection
•
Chosen algorithm can have big impact
– e.g., which filtering algorithm?
• FIR is one type, but others require less computation at
expense of lower-quality filtering
•
Example: Quickly find item’s address in 256-word
memory
– One use: data compression. Many others.
– Algorithm 1: “Linear search”
• Compare item with M[0], then M[1], M[2], ...
• 256 comparisons worst case
0:
1:
2:
3:
Linear
search
0x00000000
0x00000001
0x0000000F
0x000000FF
64
96:
128:
96
0x00000F0A
0x0000FFAA
128
Binary
search
– Algorithm 2: “Binary search” (sort memory first)
• Start considering entire memory range
–
–
–
–
If M[mid]>item, consider lower half of M
If M[mid]<item, consider upper half of M
Repeat on new smaller range
Dividing range by 2 each step; at most 8 such divisions
• Only 8 comparisons in worst case
•
a
255:
0xFFFF0000
256x32 memory
Choice of algorithm has tremendous impact
– Far more impact than say choice of comparator type
Digital Design
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Frank Vahid
94
Power Optimization
Until now, we’ve focused on size and delay
Power is another important design criteria
– Measured in Watts (energy/second)
• Rate at which energy is consumed
•
Increasingly important as more transistors fit on a
chip
– Power not scaling down at same rate as size
• Means more heat per unit area – cooling is difficult
• Coupled with battery’s not improving at same rate
8
energy (1=value in 2001)
•
•
energy
demand
4
battery energy
density
2
1
2001
03
05
07
09
– Means battery can’t supply chip’s power for as long
– CMOS technology: Switching a wire from 0 to 1
consumes power (known as dynamic power)
• P = k * CV2f
– k: constant; C: capacitance of wires; V: voltage; f: switching
frequency
• Power reduction methods
– Reduce voltage: But slower, and there’s a limit
– What else?
Digital Design
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Frank Vahid
95
Power Optimization using Clock Gating
2
•
P = k * CV f
X
•
Much of a chip’s switching f (>30%)
due to clock signals
x_ld
– After all, clock goes to every register
– Portion of FIR filter shown on right
• Notice clock signals n1, n2, n3, n4
•
•
Note: Advanced method, usually done
by tools, not designers
– Putting gates on clock wires creates
variations in clock signal (clock skew);
must be done with great care
Digital Design
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Frank Vahid
Greatly reduced
switching – less power
xt1
c1
xt2
c2
yreg
clk
n1
n2
n3
n4
y_ld
Much
switching
on clock
wires
clk
Solution: Disable clock switching to
registers unused in a particular state
– Achieve using AND gates
– FSM only sets 2nd input to AND gate to
1 in those states during which register
gets loaded
c0
xt0
n1, n2, n3
n4
a
c0
xt0
X
xt1
c1
xt2
c2
x_ld
yreg
s1
n1
n2
n3
n4
clk
s5
y_ld
clk
n1, n2, n3
n4
96
• Another method: Use low-power gates
– Multiple versions of gates may exist
• Fast/high-power, and slow/low-power, versions
size
Power Optimization using Low-Power Gates on
Non-Critical Paths
er
w
– Use slow/low-power gates on non-critical paths o
p
high-power gates
low-power gates
on noncritical path
low-power
gates
• Reduces power, without increasing delay
delay
Digital Design
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Frank Vahid
97